Installing and calibrating a complex RF system such as a satellite earth station or a cable television network requires a test equipment capable of measuring various RF signal parameters (e.g., the frequency and amplitude of RF signals).
Spectrum analyzers have long been used in the laboratory and in well equipped test facilities for analyzing RF signals. Spectrum analyzers are extremely useful because they display or plot amplitudes of significant RF signal components as a function of frequency. A spectrum analyzer typically provides a two-dimensional display (e.g., on a CRT or LCD type graphical display device) wherein the ordinate (y) axis corresponds to amplitude of RF signals and the abscissa (x) axis corresponds to frequency. FIG. 1 is a typical exemplary spectrum analyzer display output. By looking at such a display, a technician can almost instantly determine whether desired signals are missing or have abnormal amplitudes, and can also adjust the system to change relative or absolute amplitudes of certain signals. For these and other reasons, a spectrum analyzer is potentially one of the most useful pieces of test equipment in a microwave technician's toolbox.
Unfortunately, RF spectrum analyzers capable of operating at high (e.g., microwave) frequencies tend to be extremely costly. Modern digital electronics and state of the art solid state microwave components (many of which are hybrid and include digital as well as analog circuitry) have paved the way for the development of digital UHF equipment (including spectrum analyzers) that is sufficiently inexpensive and portable to be available even to field technicians. For example, so-called "sampling" type spectrum analyzers have been developed which operate by sampling signal amplitudes at discrete frequencies. U.S. Pat. No. 4,685,065 to Braun et al describes a portable sampling microcontroller-based digitally controlled spectrum analyzer providing a sweep mode in which signal amplitudes are displayed on an LCD display in bar graph form. Although such "sampling" type spectrum analyzers are particularly suited to digital control and other circuitry, they are still relatively expensive (e.g., a few thousand dollars each).
Cost continues to be a highly significant factor in marketing spectrum analyzers to people who need them. A few thousand dollars is generally too much to spend in the satellite and cable television industry to equip the average field technician with a single piece of equipment (no matter how useful it may be to him or her). Therefore, there is high demand for further improvements in sampling type digital RF spectrum analyzers that might decrease cost and complexity. Ideally, a portable RF spectrum analyzer should cost in the neighborhood of no more than about a thousand dollars or so, since this cost would not be prohibitive for field technician equipment.
Unfortunately, there are significant problems involved in providing a sufficiently accurate yet sufficiently inexpensive broadband VHF or UHF RF spectrum analyzer. These problems have caused prior art spectrum analyzers to be either too expensive, inaccurate, or both.
Since a spectrum analyzer displays RF signal amplitude as a function of frequency, accuracy considerations typically center around two RF factors: (a) the accuracy of the spectrum analyzer RF frequency selection (tuning) function, and (b) the accuracy of the spectrum analyzer amplitude detection function.
Frequency control errors can cause the spectrum analyzer to tune to frequencies other than those that are specified or desired. This can cause the spectrum analyzer to fail to detect some RF signals altogether, or to detect signals in a faulty or incomplete manner (for example, a mistuned spectrum analyzer might not center a certain RF frequency of interest within the analyzer passband, and thus might indicate an incorrect signal amplitude. Amplitude detection errors can also cause the spectrum analyzer display to misrepresent the amplitude of signal components, thus substantially decreasing the value of the information provided by the analyzer.
In some RF applications it is possible to decrease circuit cost and complexity while maximizing performance by optimizing a circuit arrangement for a particular frequency of interest. But since spectrum analyzers by definition must operate across a range or band of frequencies (and spectrum analyzer users may want this band to be wide, it is generally not possible to optimize "front end" and/or detection circuitry for a specific RF frequency. Rather, the spectrum analyzer circuitry must be capable of performing accurately across the entire frequency range of interest. It may be important for a spectrum analyzer to operate across a relatively wide frequency range e.g., the 950 MHz to 1450 MHz down-converted standard satellite television band available at the input of a standard satellite television receiver). Unfortunately, most microwave receivers are usable to simultaneously fulfill the requirements of (a) ultra high frequency operation, (b) wide bandwidth, (c) low frequency and amplitude error, and (d) low cost.
Thus, for example, low cost microwave RF amplifier stages capable of operating across a wide range of UHF frequencies typically have poor (i.e., not very "flat",) frequency response characteristics (i.e., they amplify signals at some frequencies more than they amplify signals at other frequencies, generating amplitude variations at their outputs that have nothing to do within actual input signal amplitude but which instead depend upon the idiosyncrasies of the amplifier stages themselves. Some such amplitude variations across the band spectrum are inherently created in any RF front end. In the case of inexpensive front end circuits, however, passband ripple/tilt would, if uncorrected, be reflected in amplitude errors produced on the spectrum analyzer display. Passband amplitude non-linearity can be reduced by using circuit arrangements carefully designed to provided good linearity, but such circuit arrangements are generally cost-prohibitive in the context of a low-cost hand-held spectrum analyzer.
Although digitally controlled PLL (phase locked loop) frequency synthesizer technology has advanced to a point where PLLs commonly provide highly accurate UHF frequency outputs across a wide frequency range, even this technology cannot meet to the demands placed upon the local oscillator of a UHF broadband sampling spectrum analyzer. A sampling spectrum analyzer must rapidly sample many discrete frequencies access a wide band in order to "sweep" across the frequency range of interest.
In PLL design it is generally necessary to trade off loop response time for wide bandwidth with a large number of frequency increments. A UHF spectrum analyzer must provide relatively narrowly spaced frequency increments (e.g., 5 MHz) over a relatively wide frequency range (e.g., on the order of 500 MHz). At these frequencies, a PLL used to control the VCO must provide a rather large "divide-by-N" factor (the local oscillator frequency must be relatively low in order to provide requisite frequency base stability and resolution) causing dramatic increase of the PLL lock time. It is generally not possible to provides a phase locked loop that can rapidly lock on to a large number of frequency points over a wide frequency range; rapid loop lock times are generally achievable only in loops having small operating frequency ranges and/or a small number of controllable frequency increments. Unfortunately, it is necessary in a spectrum analyzer to sweep the local oscillator frequency across the wide spectrum at relatively rapid rate (e.g. 4 full sweeps per second) -- and the PLL VCO must thus provide as many as 400 frequency changes per second.
This serious shortcoming in PLL performance has in the past required either slower scan times, fewer frequency sampling points, or increased frequency synthesizer complexity (e.g., use of multiple PLLs with their outputs mixed together).
Moreover, inexpensive components typically offer decreased precision and accuracy and may also tend to drift in response to temperature or other variations. For example, inexpensive digital-to-analog converters typically offer poor precision and accuracy; yet, such converters providing requisite precision and accuracy may be too expensive for use in a spectrum analyzer that is to sell for no more than a thousand dollars.
The spectrum analyzer described in the Braun patent does not solve the problems mentioned above. This system is generally too expensive and does not provide UHF (e.g., on the order of 1000 MHz) operating range. Braun's system provides a VCO which is operated under PLL control--thus exhibiting seriously limiting the scanning rate and/or number of samples across a wide band. The Braun patent discloses that "the microprocessor maintains proper [amplitude] calibration of the instrument by turning off the input signal, switching on the noise source 5, and taking level readings." See column 8, lines 37-39. However, Braun does not disclose any way to provide accurate automatic amplitude calibration using inexpensive RF components.
Danzelsen U.S. Pat. No. 4,758,783 discloses a spectrum analyzer that includes a digital memory used to store frequency dependent amplitude control values. The memory contents are converted to analog form by a DAC and used to control the gain of a variable gain I amplifier 5. Memory entries are selected by converting the output of a sweep generator oscillator to digital values, and using the digital values to address the memory. The Danzelsen patent fails to disclose any automatic calibration technique for automatically storing values into the memory or for automatically generating such stored values which take DAC non-linearity into account.
U.S. Pat. No. 3,681,577 to Gasiunas teaches a digital gain correction/calibration arrangement for a multiple channel atomic spectroscopy analyzer.
U.S. Pat. No. 4,652,816 to Crookshanks discloses a spectrum analyzer having a memory providing digital calibration phase and amplitude control obtained by network analysis.
U.S. Pat. Nos. 4,430,611 and 4,672,308 disclose spectrum analyzers providing various techniques for reducing frequency errors.
The following additional patents are generally relevant to instrumentation calibration techniques:
U.S. Pat. No. 3,988,667 to Roth et al;
U.S. Pat. No. 4,162,531 to Rode et al;
U.S. Pat. No. 4,198,677 to Brunner et al;
U.S. Pat. No. 4,200,933 to Nickel et al;
U.S. Pat. No. 4,270,177 to Endoh et al;
U.S. Pat. No. 4,377,517 to Nickel et al;
U.S. Pat. No. 4,453,218 to Sperinde et al;
U.S. Pat. No. 4,509,132 to Kavaya;
U.S. Pat. No. 4,542,638 to Tlaker;
U.S. Pat. No. 4,573,133 to White;
U.S. Pat. No. 4,633,173 to Kashiwagi;
U.S. Pat. No. 4,667,151 to Crookshanks;
U.S. Pat. No. 4,703,433 to Sharrit;
U.S. Pat. No. 4,714,929 to Davidson;
U.S. Pat. No. 4,733,234 to Sparks et al;
U.S. Pat. No. 4,801,861 to Ashley et al;
U.S. Pat. No. 4,858,159 to Wheelwright; and
U.S. Pat. No. 4,918,373 to Newberg.
Of these listed patents, the White patent (U.S. Pat. No. 4,573,133) discloses digital map gain correction techniques; Sharrit U.S. Pat. No. 4,703,433 discloses a complex microwave frequency "network analyzer" including Fourier analysis capabilities; Rode et al (U.S. Pat. No. 4,162,531) discloses digital gain adjustment circuitry; and Crookshanks (U.S. Pat. No. 4,667,151) teaches a digital memory controlled RF sweep generator.
None of the prior art described above provides an RF spectrum analyzer that is sufficiently cost effective and accurate and is also capable of operating over a relatively wide RF range.
The present invention, in contrast, provides a hand-held portable low cost frequency spectrum analyzer designed to operate over a wide RF band (e.g., in the so-called "L band" range of 950-1450 MHz). The spectrum analyzer provided by the present invention uses low cost, relatively uncomplicated components and circuit arrangements, but is able to automatically calibrate and compensate for amplitude and frequency errors arising from such components and arrangements. The spectrum analyzer of the present invention thus provides low cost as well as good accuracy over a wide frequency range.
In accordance with one aspect of the present invention there is provided an arrangement and associated technique for calibrating a microcontroller-based spectrum analyzer to overcome the amplitude non-linearity passband problems. The analyzer is automatically calibrated (e.g., at the factory) for amplitude variations under software control. Resulting software-defined calibration parameters are used during system operation to reduce or eliminate passband amplitude errors.
Briefly, a microcontroller in the preferred embodiment provided by the present invention is coupled to a DAC (digital to analog converter) the output of which is used to control the output current (Is) of a current source. The current source output controls the gain of a variable gain intermediate frequency (IF) amplifier.
During factory calibration, a broadband noise spectrum of known constant (over frequency) amplitude is applied to the input of the spectrum analyzer. The microcontroller executes a software routine causing it to step the system incrementally through frequencies in the spectrum of interest and to measure the signal amplitude provided at the output of the IF stages for each incremental frequency. The microcontroller compares the measured signal amplitude with a standard (e.g., software specified) signal amplitude, and adjusts the gain of the variable gain amplifier until the result of the comparison (i.e., the error signal) is zero. The microcontroller stores the associated digital control value applied to the DAC. This stored value is inherently corrected for non-linearity in the DAC, the IF variable gain amplifier, and the current source. The value is stored in non-volatile memory in a gain correction table entry corresponding to the incremental frequency. The analyzer then moves on to a next frequency. At the end of this calibration procedure, the microcontroller has constructed a table of gain correction factors for different operating frequencies.
During normal operation of the spectrum analyzer, the gain correction factor table is accessed each time the spectrum analyzer tunes to a new frequency. The microcontroller applies the amplitude correction factor corresponding to each new frequency (interpolating from gain correction table entries for operating frequencies which do not correspond to table entries) to the DAC so as to adjust the gain of the IF amplifier to correspond to a gain exhibiting a zero error characteristic for that frequency. In this way, amplitude variations due to front end attenuator, IF and detector non-linearities are substantially eliminated.
In accordance with another aspect of the present invention, a phase locked loop is closed to form a closed loop only during calibration (when DAC control values are obtained and stored). During normal operation, the PLL VCO is connected in an open loop circuit with a DAC. This technique of using the PLL in closed loop connection during calibration and operating the VCO in open loop connection in response to values derived by software during calibration achieves nearly the accuracy of closed loop operation while avoiding the penalty of long PLL lock time.
In somewhat more detail, the preferred embodiment uses low-cost DACs (digital-to-analog converters) to control the VCO in an open loop arrangement--even though the low cost DACs themselves introduce additional problems of low resolution and poor linearity. A software calibration routine (and associated components) are provided to calibrate the spectrum analyzer so as to eliminate frequency errors. This calibration routine is performed by the preferred embodiment each time power is initially applied to the spectrum analyzer (and in the preferred embodiment is performed continually between analyzer sweeps to provide on-going automatic calibration).
The microcontroller periodically steps the VCO incrementally through the tuning range using PLL control (i.e., waiting a sufficient time to permit the PLL loop to settle and lock onto each new frequency). For each frequency, an A/D converter coupled to the VCO input samples the VCO input control voltage provided by the loop to the VCO input for each such locked frequency step, and the microcontroller stores the resulting digital value corresponding to the measured control voltage.
After the PLL settles and its phase comparator output is measured for each frequency, the microcontroller applies the stored (possibly scaled) value obtained from the measured VCO control voltage to a DAC. The microcontroller also disconnects the VCO from PLL (phase comparator) control and connects the VCO control voltage input to the output of the DAC. At the same time, the microcontroller monitors the PLL phase comparator output (the PLL still being programmed for the frequency and connected to receive the VCO output), and uses the open loop PLL as a frequency comparator. The microcontroller determines from the PLL phase comparator output whether there is (and the sign of any) error in the frequency output now produced by the VCO under DAC open loop control. The microcontroller adjusts the digital value applied to the DAC (using an iterative binary approximation routine in the preferred embodiment) so as to minimize error between the VCO control voltage applied to the VCO by the PLL and the VCO control voltage applied to the VCO by the DAC.
This process is performed for several frequencies across the spectrum (to reduce "power on" calibration time, the process is not performed for all possible frequency increments in the preferred embodiment, and only a limited number of iterations are performed for each frequency). The resulting frequency calibration (i.e., DAC control) values are recorded in a frequency calibration table stored in memory. During normal operation of the spectrum analyzer, the VCO is operated in an open loop connection and receives its control voltage from the DAC. The microcontroller controls the control voltage provided at the DAC output in response to the values stored in the frequency calibration table, interpolating between stored values where necessary. Meanwhile, the microcontroller continually performs the calibration technique in a piecemeal fashion for as long as the spectrum analyzer is operating (e.g., by adjusting frequency calibration table values between spectrum sweeps, using the PLL in open loop connection as a frequency comparator) so as achieve increased calibration resolution without increasing power on calibration time (and to also permit the analyzer to adapt to changing circuit parameters and drift).
The amplitude and frequency calibration techniques and arrangements discussed above permit the spectrum analyzer to have relatively inexpensive components and circuit arrangements and nevertheless achieve high accuracy. Using the techniques and arrangements provided by the present invention, it becomes possible to provide a spectrum analyzer that is much less expensive than prior art spectrum analyzers which provides accuracy sufficient for most field requirements.